Interspersed training for turbo coded modulation

ABSTRACT

A communications system, having a combination Reed-Solomon encoder and a Turbo-Code encoder Data frame configuration which may be changed to accommodate embedded submarkers of known value. The submarkers are embedded in with the data order to aid synchronization in the receiver system, by providing strings of known symbols. The string of known symbols may be the same as the symbols within a training header that appears at the beginning of a data frame. Frame parameters may be tailored to individual users and may be controlled by information pertaining to receivers, such as bit error rate of the receiver. Additional headers may be interspersed within the data in order to assist in receiver synchronization. Frames of data may be acquired quickly by a receiver by having a string of symbols representing the phase offset between successive header symbols in the header training sequence in order to determine the carrier offset. Phase lock to a signal may be achieved after determining carrier offset in receivers by correlating successive symbols in successive headers.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.10/822,403, filed Apr. 12, 2004 now U.S. Pat. No. 6,828,926 entitledINTERSPERSED TRAINING FOR TURBO CODED MODULATION, which is acontinuation of U.S. application Ser. No. 10/703,286 entitledINTERSPERSED TRAINING FOR TURBO CODED MODULATION filed on Nov. 7, 2003,now U.S. Pat. No. 6,891,485 which is a divisional of patent applicationSer. No. 09/729,442 entitled INTERSPERSED TRAINING FOR TURBO CODEDMODULATION filed on Dec. 4, 2000 (now U.S. Pat. No. 6,693,566) whichclaims priority from provisional application No. 60/168,808 entitledINTERSPERSED TRAINING FOR TURBO CODED MODULATION filed on Dec. 3, 1999,the contents of which is expressly incorporated by reference as thoughset forth in full.

FIELD OF THE INVENTION

The present disclosure relates to digital signal reception and, inparticular, signal coding which assists in the synchronization ofreceivers with turbo decode capability.

BACKGROUND OF THE INVENTION

In recent years, transmission of data via satellite has increasedconsiderably. Recently, the number of personal satellite receivers havealso been increasing. As large satellite receiving antennas andexpensive receivers are replaced by smaller and less expensiveequipment, the demand for such systems continues to rise. As the demandfor satellite communication systems rises, systems which have increasedperformance have a distinct market advantage. Improving designs andincreasing the level of system integration within satellite receiverscan offer the dual benefits of decreasing system costs and increasingperformance. Accordingly, there is a need for improved satellitecommunication systems within the art.

SUMMARY OF THE INVENTION

In one aspect of the present invention, a method of creating a datasequence includes placing a training sequence at a beginning of a dataframe, placing a plurality of the blocks of turbo encoded data withinthe data frame following the training sequence, and interspersing aplurality of submarkers within the turbo encoded data blocks.

In another aspect of the present invention, a training sequence andsubmarker insertion apparatus includes an input adapted to receive aplurality of turbo encoded data blocks, and an inserter adapted toinsert a training sequence before the turbo encoded data blocks andinsert a plurality of submarkers within the turbo encoded data blocksthereby creating a data frame.

In yet another aspect of the present invention, a training sequence andsubmarker insertion apparatus includes receiving means for receiving aplurality of turbo encoded data blocks, insertion means for inserting atraining sequence before the turbo encoded data blocks, and inserting aplurality of submarkers within the turbo encoded data blocks therebycreating a data frame.

In a further aspect of the present invention, a transmitter includes aforward error correction device having a turbo encoder and a trainingsequence and submarker insertion device coupled to the turbo decoder,the training sequence and submarker insertion device comprising an inputadapted to receive a plurality of turbo encoded data blocks from theturbo encoder, and an inserter adapted to insert a training sequencebefore the turbo encoded data blocks and insert a plurality ofsubmarkers within the turbo encoded data blocks thereby creating a dataframe, and a modulator coupled to the forward error correction device tomodulate the data frame.

In yet a further aspect of the present invention, a method of creating adata sequence includes turbo encoding data into a plurality of turboencoded data blocks, creating a data frame comprising a first portionand a second portion, the first portion preceding the second portion intime, placing a training sequence in the first portion of the dataframe, placing the turbo encoded data blocks in the second portion ofthe data frame, and interspersing a plurality of submarkers within theturbo encoded data blocks.

It is understood that other embodiments of the present invention willbecome readily apparent to those skilled in the art from the followingdetailed description, wherein it is shown and described only embodimentsof the invention by way of illustration of the best modes contemplatedfor carrying out the invention. As will be realized, the invention iscapable of other and different embodiments and its several details arecapable of modification in various other respects, all without departingfrom the spirit and scope of the present invention. Accordingly, thedrawings and detailed description are to be regarded as illustrative innature and not as restrictive.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features, aspects, and advantages of the presentinvention will become better understood with regard to the followingdescription, appended claims, and accompanying drawings where:

FIG. 1 is a graphical representation of an example environment in whichembodiments of the present invention may operate.

FIG. 2 is a graphical illustration of a data format as may be used withembodiments of the present invention.

FIG. 3 is a graphical illustration of frequency versus amplitude graphsrepresenting a band of signals with different center frequencies(carrier offsets).

FIG. 4 is a graphical representation of header symbols being transmittedthrough a communications channel.

FIG. 5 is graphical illustration of a mechanism that may be used tosearch for given sequence of symbols, such as those found in a header.

FIG. 6 is a graph illustrating correlation values versus frequencyutilizing a system for a short sequence illustrated in FIG. 5.

FIG. 7 is a graph of correlation value versus frequency when a longsequence is being correlated.

FIG. 8 is a graphical illustration of a step in the process, which canfind the proper demodulation frequency of a signal, without thenecessity of a series of cut and try steps.

FIG. 9 is a graphical representation of a differential correlator.

FIG. 10 is a graphical illustration relating header data to differentialcorrelator output and further relating header position to a counter,which counts cycles of a receiver clock.

FIG. 11 is a graphical illustration of the process by which thefrequency of the receiver clock may be synchronized to an incoming datastream, using a clock counter.

FIG. 12 is a block diagram illustrating the parts of a common correlatoras may be used to establish phase lock to a received signal.

FIG. 13 is a graphical illustration of a serial correlator as may beused in embodiments of the present invention.

FIG. 14 is a block diagram representation of subsystem circuitry, whichcomprises the modulator and forward error correcting sections of atransmitter system according to an embodiment of the invention.

FIG. 15 is a block diagram illustrating the component and signal flowwhich comprise an exemplary embodiment of forward error correctionaccording to an embodiment of the invention.

FIG. 16 is a graphical illustration of a generalized data frame,according to an embodiment of the invention.

FIG. 17 is a graphical illustration of a frame of data having threesubmarkers disposed therein.

FIG. 18 is a graphical illustration of an exemplary arrangement of datablocks and submarkers within a frame.

FIG. 19 is a block diagram of a set top box according to an embodimentof the current invention.

FIG. 20 is a block diagram of a receiver according to an embodiment ofthe invention.

DETAILED DESCRIPTION Exemplary Communication System

In the following description, reference is made to the accompanyingdrawings, which form a part hereof, and in which is shown, by way ofillustration, specific embodiments illustrating ways in which theinvention may be practiced. It is to be understood that otherembodiments may be realized, as the inventive concepts disclosed hereinmay be used in the design and fabrication of other embodiments, withoutdeparting from the scope and spirit of the inventive concepts disclosedherein.

Accordingly, embodiments of the present invention relate, generally, tosatellite communication systems. However, for the purposes ofsimplifying this disclosure, the embodiments are described herein withrelation to direct broadcast satellite systems. Although the describedexemplary embodiments disclosed herein are directed to direct broadcastsatellite systems, there is no intent to limit the invention to theexample embodiments. The exemplary embodiments are intended toillustrate inventive aspects of the present invention, which areapplicable to a wide variety of electronic systems.

Satellite communication systems in general comprise three parts. Thefirst part is a transmit system also known as a ground station. Thetransmit system may receive data from a variety of sources such as cablecompanies, Internet providers, etc. The received data is then coded,modulated and provided to a transmitter for broadcast. Coding generallyincludes a forward error correcting code. A forward error correction andmodulation section of a transmitter accepts digital data and thenconstructs a signal that will be relayed, via a satellite, to the user.The second element in the satellite communication system is a satellite,which is illustratively a geo-synchronous satellite. The third part ofthe satellite communication system is a receiver system. The receiversystem generally comprises an outdoor unit and a receiver which receivesthe signal from the outdoor unit, demodulates it and decodes it torecover the original signal. The original signal is then available foruse in a device such as a television or computer.

FIG. 1 is a graphical representation of an example environment, in whichthe described exemplary satellite communication system may operate.Within the environment of FIG. 1, a data source 101 provides data to aground station 100, which then broadcasts the data to a satellite 109.The satellite in turn re-broadcasts the data to a receiver system 110thereby providing it to a user device 117.

In FIG. 1 the data source 101, such as a television cable signal,represents an example of one of a number of various types of datasignals, which may be conveyed by the system. For example, the datasignals may include, but are not limited to, television channels, musicchannels, or data from Internet websites. The data signals are provided,by the data source 101, to a modulator and forward error correction(FEC) 103. Within the modulator and FEC 103, the data is modulated onone or more carrier waveforms. The modulator and FEC 103 translates thedata from data source 101 into a form suitable for transmission. Themodulated data is further coupled to a transmitter 105. In thetransmitter 105, the data stream from the modulator and FEC may befurther amplified and coupled, for example, to a dish antenna 107 fortransmission to the satellite 109.

The satellite 109 accepts data transmitted from dish antenna 107.Satellite 109 is commonly a geo-synchronous satellite, in which thesatellite's position is a constant location above the earth, but is notlimited to such. In a geo-synchronous satellite, orbital rotation of thesatellite is one day exactly matching the rotational speed of the earth,thereby maintaining the satellite at a constant position above theearth. The data transmission is accepted by the satellite 109, amplifiedand rebroadcast to the receiver system 110 on earth.

A user antenna 111, accepts the data transmission from the satellite109. The received signal is then further coupled into a LNB (low noiseblock) 113 where it is amplified. The LNB 113 then provides the signalto a set top box 115. Within the set top box 115, the signal isdemodulated and converted into a form which may be used by the userdevice 117, such as a television or computer. The LNB 113 and theantenna 111 are collectively referred to as the ODU (outdoor unit) 119because they are typically located outdoors.

Transmission of Data Between the Transmit and Receiver System

FIG. 2 is a graphical illustration of a data format as may be used withembodiments of the present invention. The data stream is divided into aseries of successive concatenated frames. Three successive concatenatedframes are illustrated in FIG. 2 at 301, 303 and 309. A typical frame301 may comprise a header 305 followed by data 307. Coupling a header305 along with data 307 in a discrete frame 301 may provide particularadvantages. For example, the frame 301 may have a different modulationscheme than a successive frame 303. Frame 301 may contain QPSK symbolsand frame 303 may contain 8 PSK symbols, and each frame may be intendedfor different receivers. Additionally a different format can be used forthe header and the data, for example a QPSK header may be used with 8PSK data. The header may be encoded so as to indicate the format of thedata within the block, or of future blocks. Additionally the header maycontain a “training sequence” to assist the receiver in synchronizing tothe transmitted signal. A great variety of combinations of datapartitioning are possible. Each frame may be intended for differentusers and hence may have different types of modulation and data formatsin successive data frames. By using such a flexible scheme for datadelivery, a variety of user needs can be accommodated.

Compensating for Frequency Offsets

FIG. 3 is a graphical illustration of frequency versus amplitude graphsrepresenting a band of signals with different carrier (center)frequencies having differing carrier offsets. The band-pass curve 705represents a nominal case in which F_(c) (the desired center frequency)is situated in the center of the bandwidth of the signal 705. In actualpractice, the frequency band may be offset from the desired center asillustrated in bandwidth curves 707 and 709. This offset may be due to avariety of factors, such as drift in the transmitter or receiver frontend, drift in the relay satellite, and in non Geo-stationary systems dueto Doppler effects. In curve 707, the center frequency has beendisplaced and is actually no longer equal to F_(c) but is equal toF_(c)+ΔF. In curve 709, the carrier frequency is actually lower thanF_(c), the actual center frequency for band-pass curve 709 is located atF_(c)−ΔF. In practice, offsets can cause considerable displacement of asignal's center frequency. Offsets in carrier frequency typically mustbe accounted for to assure proper reception. Therefore, finding theactual carrier frequency is commonly an early step in locking a receiversystem to a transmitted signal.

In an exemplary embodiment of the present invention, header data isidentified and, in the process of identifying the header data, thefrequency offset of the communications channel carrier frequency isfound. One method for determining the carrier frequency offset at thereceiver system is to mix the incoming signal with series of frequenciesone at a time until the correct frequency is found. By mixing theincoming signal across a series of frequencies and correlating theresultant signal (for example a resulting baseband signal), the actualcenter frequency of the incoming signal can be determined. Mixing theincoming signal with a series of frequencies, however, can takeconsiderable amount of time. The number of frequency offsets that mayhave to be applied to the incoming signal, before the correct frequencyoffset is found, can be considerable. It is desirable to be able todetermine the offset of the center frequency without going through aprocess of trial and error, incrementing the mixing frequency and usingthe incremented frequency to mix with the incoming signal.

FIG. 4 is a graphical representation of header symbols being transmittedthrough a communications channel. A sequence of symbols 801 isrepresented by A_(N). The sequence A_(N) contains a number of symbolsH_(L) which is equal to the header length. A_(N) represents a sequenceof header symbols for N=0 to N=(H_(L)−1).

The header symbols A_(n) are coupled into communications channel 803.The symbols are then accepted from the communication channel by areceiver system. The received symbols R_(N) form a set of symbols 805that have had noise and/or distortion added to them as a consequence oftransmission through the communications channel. The sequence of symbolsR_(N) can be represented by a modeling mathematical expression A_(N)e^(−jwt)+N. Term N (809) represents the noise added by thecommunications channel and term e^(−jwt) (807) is a frequency component,representing the frequency offset of R_(N)'s carrier. The communicationschannel 803 generally comprises everything between the transmitter andthe user's receiver system.

For a given frequency offset represented by term 807, H_(L) (the lengthof the sequence A_(N)) determines how large of a frequency offset can betolerated and still form a proper correlation. The longer the sequenceof symbols that are being detected in a correlator, the less frequencyoffset that can be tolerated in the center frequency. Therefore, inorder to tolerate a large frequency offset, a short sequence isdesirable. On the other hand, a long sequence will result in more gainwhen correlated with the sequence to be detected. So from the standpointof correlator gain, a long sequence is desirable. Generally, correlatorgain is more important than being able to tolerate a large frequencyoffset, because the correlator gain may be used to offset the noisewithin the channel.

FIG. 5 is a graphical illustration of a mechanism in the receiver systemthat may be used to search for a given sequence of symbols, such as“training symbol” found in a header. Training symbols found in theheader are known symbols, which a receiver system may look for in orderto synchronize or “lock to” a received transmission. The output of thecommunications channel 803 is provided to a demodulator 903 in thereceiver system. The demodulation frequency F_(D) 901 is also coupledinto the demodulator 903. It is F_(D) 901 that will be mixed with theincoming signal and that must be properly adjusted to translate theincoming data symbol stream, from the communications channel 803, into abaseband sequence of received symbols R_(N), 805. The baseband symbolstream R_(N), (805), is coupled into a correlator 905. The correlator905 will search for a match between the header sequence A_(N) and thereceived symbols R_(N). The correlator is clocked by a clock 905 whichcontrols the comparison between the known header A_(N) and the receivedsymbol stream R_(N). The comparison of A_(N) and R_(N) within thecorrelator 905 is can be a bit by bit comparison. The bits that matcheach other are typically added to produce a correlation value 919. Thecorrelation process can be used to ascertain the center frequency of thecarrier of the received symbol R_(N) data stream. For example, F_(D) 911may be changed in steps (swept) and the correlation value, whichresults, saved. Once the demodulation frequency has been swept acrossthe range of possible values, the correct demodulation frequency can beascertained by observing which frequency step gives the highestcorrelation value.

FIG. 6 is a graph illustrating the correlation value for a shortcorrelation sequence versus frequency for the correlation systemillustrated in FIG. 5. The maximum correlation value is given by point1009. The correlator output 1005 tends to have a wider curve as thesequence correlated becomes shorter. However, the maximum output 1009tends to become smaller as the sequence becomes shorter, thus yielding alower gain.

FIG. 7 is a graph of correlation versus frequency offset for a longsequence being correlated by a system such as illustrated in FIG. 5.When a long sequence is correlated, the curve produced 1103 is muchnarrower and its peak value is higher than a short sequence. This meansthat the correlator has more gain, but that finding F_(D) 901 isgenerally of greater concern. In other words, a shorter sequence willallow larger steps in F_(D) 901 to be used in finding the correct F_(D),than will a longer sequence. Therefore, to search for a longer sequence,the frequency steps that are used as the demodulator frequency 901should be closer in value and, hence, may comprise more steps than acorresponding shorter sequence. Even if a shorter correlation sequencecan be tolerated, the process of trial and error in order to determinethe correct frequency of demodulation 901 can be cumbersome andtime-consuming. It is advantageous to eliminate the trial and errormethod of determining the necessary demodulation frequency.

FIG. 8 is a graphical illustration of a step in the process, which canfind the proper demodulation frequency of a signal without the necessityof a series of trial and error steps. Equation 1201 represents a seriesof N header symbols. The series of header symbols are part of a dataframe, for example 301, as illustrated in FIG. 2. The satellitetransmission may contain a header 305 and data 307 in a frame 301 as aconvenient way of packaging satellite transmissions. The header may beknown to a receiver system beforehand and hence is a convenient datasynchronization pattern. A known header, called a training sequence, isused to adjust the receiver system to receive the data frame.

From the header symbols, as given in equation 1201, a secondary sequence1203 can be created. The secondary sequence, represented by B_(N) is asequence of symbols which represents the phase difference betweensuccessive symbols within the header. So, for example, the symbols inthe sequence B_(N) are found by taking the phase difference betweenA_(N) and A_(N−1). Symbol B_(N−1) is formed by taking the differencebetween symbol A_(N−1) and A_(N−2). The remaining symbols in the B_(N)sequence are formed similarly, as illustrated in equation 1203. Thegeneral formula for the B_(N) sequence is given in equation 1205.Because the header A_(N) is known beforehand by the receiver system, thesequence B_(N) also may be known beforehand (or computed) by thereceiver system.

A sequence S_(N) is formed by taking the phase difference betweensuccessive received symbols R_(N). This sequence is as illustrated inequation 1207.

FIG. 9 is a graphical representation of a differential correlator in thereceiver system that can be used to correlate the S_(N) and B_(N)sequences of symbols. The differential correlator 1305 correlates thephase differential between successive symbols. B_(N) is the sequence towhich the received symbol differential sequence S_(N), 1303, iscompared. [The sequence B_(N) may be time reversed and a conjugatedversion of B_(N) (B_(N)*)]. When the received symbol differential stringS_(N) is compared to the known differential sequence B_(N), a series ofheader peaks 1307 are produced at the output of the differentialcorrelator 1305.

FIG. 10 is a graphical illustration relating the header data to thedifferential correlator output and further relating header timing to acounter, which counts cycles of a receiver system clock. Thedifferential correlator 1305 receives the differential symbol streamS_(N). The differential correlator 1305 also has preloaded the B*_(N)sequence, 1301. The correlation between output 1415 of the differentialcorrelator 1305 compared to the received symbol sequence R_(N) is shownin graph 1419. Once a complete header has been received and thedifferential sequence S_(N) created from the header, a differentialcorrelator peak 1401 will be seen from the correlation between S_(N) andB_(N).

Concurrently with the operation of the differential correlator 1305, aclock counter 1407 counts cycles of the receiver system clock 1413. Theclock counter 1407 is configured so that it rolls over, that is, oncethe clock counter 1407 achieves a maximum value, the counter isautomatically reset to zero upon the receipt of the next clock pulse.The clock counter 1407 is configured so that its maximum count willcorrespond to a period that is longer than the combined period of theheader and data. When the clock counter 1407 is free-running, i.e.,counting receiver system clock cycles and resetting upon a maximum clockcounter count, its output period is guaranteed to be longer than thereceived header plus data packet. Because the roll over period of theclock counter is longer than the longest combined period of a headerplus data, it is guaranteed that, between rollovers of the clockcounter, a maximum differential correlator value, for example 1401, willoccur. Because a maximum value is being determined, the system is notdependent on a particular threshold value. Because there is always amaximum value within the clock counter period, any design difficultiesencountered by attempting to adjust a correlator output threshold forchanging signal conditions is eliminated. The output counter value 1417is portrayed as a saw-toothed waveform. This portrayal is for purposesof illustration and comparison only. The output 1417 of clock counter1407 is generally a sequence of increasing integers, and therefore, theoutput counter value 1417, instead of being a sawtooth as portrayed, isgenerally a series of discrete steps.

Once the maximum differential correlator value has been ascertained forthe free-running clock counting period, the clock counter output valuecorresponding to the differential correlator peak value within the clockcounter period can be determined. The output counter values 1403 and1409 correspond to maximum differential correlator values. By knowingthe values of the clock counter that correspond to successivedifferential correlator peaks, i.e. 1401 and 1409, the actual period ofthe header plus data can be determined.

FIG. 11 is a further graphical illustration of the process by which thereceiver system clock may be synchronized to the incoming data streamusing the clock counter 1413. Curve 1501 represents the output of thedifferential correlator 1305. Initially, the receiver system clockcounter 1413 is in a condition of free-run, as illustrated in curve1503. When the clock counter 1413 is free-running, it will count thereceiver system clock pulses up to a maximum number and then roll over.The time between roll over of the clock counter is referred to as theclock counter period, for example 1525. By observing the output of thedifferential correlator 1305 during a full clock counter period, forexample 1525, a time between peak correlator output values can bedetermined.

Once the peak correlator values 1509 and 1517 have been determined,corresponding values 1511 and 1519 of the clock counter 1413 can bedetermined. Once the values 1511 and 1519 have been determined, thedifference between them can be determined. The free-run frequency of thereceiver system clock, and hence the free-run frequency of the clockcounter 1413, which counts the cycles of the receiver system clock, canbe adjusted so that the difference between 1511 and 1519 is zero. Whenthe receiver system clock 1413 has been adjusted to a point where values1511 and 1519 are equal, the receiver system clock has been synchronizedin frequency with the transmitter clock. The frequency adjustment isaccomplished by decreasing the clock counter period, as illustrated by1527 of the waveform 1505.

Those skilled in the art will recognize that the preceding descriptionis exemplary only and many variations of this scheme are possible. Forexample, instead of merely measuring the time difference between points1511 and 1519, a series of points may be measured and an average timedifference may be computed. Similarly, two maximum correlator values canbe selected from a number of correlator peaks in order to select, forexample, the maximum value of correlation peaks over a series ofmultiple correlation peaks. Once the frequency has been adjustedcorrectly; (as illustrated in wave form 1505) the timing differencebetween 1511 and 1521, which represents the clock counter value, atwhich correlation peaks 1509 and 1517 occur, should be zero.

Once the frequency of the receiver system clock counter 1413 has beenadjusted to match the transmit clock, the receiver system clock can thenbe adjusted to match the phase of the transmit system. Because thedifferential correlator 1305 correlates the phase difference betweenadjacent symbols, it is essentially phase blind. That is, once asequence has been created, such as S_(N) or B_(N), which comprises aphase differential between symbols, the actual phase informationrelative to each symbol is lost. To establish phase lock with respect toincoming symbols, the received sequence is correlated in real time.

FIG. 12 is a block diagram illustrating the parts of an exemplarycorrelator, as may be used to establish phase lock. An input sequence1601 is coupled into a correlator 1603. The input sequence 1601 is thencoupled into a tapped delay line 1605. The stages of the tapped delayline are compared in a series of comparators 1607, to a known sequence1609. The output of all the comparisons are then summed in a summationunit 1611, which produces the correlator output 1613. One drawback of acorrelator, such as illustrated at 1603, is that it is very areaintensive to fabricate such a correlator on an integrated circuit. Thatis, if the input sequence 1601 happens to be 128 symbols long, then 128multipliers, for the multiplier chain 1607, and 128 addition stageswithin the summation unit 1611 will need to be employed. It is desirableto find a method that is less area intensive when implemented on aintegrated circuit chip.

FIG. 13 is a graphical illustration of a serial correlator as may beused in certain embodiments of the present invention. A series ofcomparison patterns, such as packet headers B_(N), are illustrated. Thesequential packet headers are numbered 1701, 1703, 1705, 1707, and 1709.The pattern that will be compared to the header pattern is contained ina memory 1713. Instead of comparing all of the bits at the same time, aswith the exemplary correlator illustrated in FIG. 12, the serialcorrelator compares one bit at a time. For example, B_(O) from the firstheader 1709 is compared with a first bit C_(O), 1715, in the pattern tobe matched. The single bits are compared in a single bit multiplier1711. The result of the comparison, of the individual bits in themultiplier 1711, is provided to a summation unit 1719. So, for example,if a pattern represented by C₀ 1715 through C_(N) 1717 is being sought,the Nth bit of the header B_(N) represented by 1701 may be compared tothe C_(N) bit 1717 of the pattern to be correlated. When the next headerarrives, the C_(N−1) bit from the comparison stream 1713 can be comparedto the B_(N)−1 bit in the second header to arrive 1703. In this manner,not all the bits are compared at once. Instead of requiring, forexample, 128 multipliers and an adder chain capable of adding 128results, the serial correlator correlates one bit at a time and thensums the result. Once 128 header bits have been compared with thecorrelation value 1713 the correlator output 1721 becomes valid.Although the serial correlator can compare only one header bit at atime, the actual delay in achieving phase lock may not be perceptible toa user. By serializing the computation, a great savings in chip area maybe realized.

The Transmit System

FIG. 14 is a block diagram representation of a modulator and FEC of atransmit system according to an embodiment of the invention. The FECaccepts data 1807 from an outside source. The data may be of anysuitable type, for example, MPEG 2 (Motion Picture Experts Group) Video,DVB (Digital Video Broadcast), or PN (Pseudo-Noise) codes. In theexemplary embodiment, the FEC also provides a test data signal 1805,which can be used as a data source input 1807 to the FEC. The datasignal 1805 can be used for testing, for example, in a stand-alone mode.Because the test signal is a known data pattern, the receiver system canuse the same pattern to measure bit error rate. Various parameters ofthe circuitry may influence the bit error rate. The built in test signal1805 may reduce the necessity of having a dedicated piece of testequipment in order to provide a known test signal to measure the biterror rate.

The FEC 1801 also includes a data clock 1809. The data clock 1809 clocksthe input data 1807 and thereby controls the rate that the data isprovided to the FEC 1801. The FEC also accepts clocking signals fromseveral numerically controlled oscillator clocks. The exemplaryembodiment accepts clocking inputs from a byte clock 1811 (which mayalso function as a data clock 1809), a forward error correcting clock1813, and a bit clock 1815. The FEC also accepts a control input 1825.The control input 1825 may be used to control various parameters of theFEC, for example, but not limited to, modulation format, frame length ofthe data exiting the module, and coding rates. Once the FEC 1801 hasprocessed the data 1807, the processed data 1819 is further provided tothe modulator 1803. The modulator also communicates with the FEC via acommunications bus 1817. The communications bus 1817 generally provideshandshaking and protocol to control the data transfer over the bus 1817.The modulator 1803 also accepts an input from the control line 1825. Thecontrol line 1825 may be used to control various parameters within themodulator, such as filters, equalizers, and symbol mappers. Themodulator 1803 also accepts a clocking signal, such as a 100 megahertzoscillator 1823. The oscillator 1823 also is coupled through themodulator and operates a digital to analog converter 1827 which acceptsthe data 1827 output by the modulator 1803.

Overall, the data 1807 enters the FEC 1801. The FEC codes the data andpasses the coded data 1819 to the modulator 1803. The modulator 1803accepts the data from the FEC 1801, modulates it, and places themodulated data on an output 1827. The data output 1827 is then coupledinto a digital to analog converter. The digital to analog converterconverts the digital signal 1827 into an analog signal suitable to beaccepted by the transmitter 105 and broadcast by the antenna 107 to thesatellite 109 (see FIG. 1).

Forward Error Correction

FIG. 15 is a block diagram illustrating the components and signal flowwhich comprise an exemplary embodiment of the FEC 1801. The FEC isdivided into three asynchronous modules. The first asynchronous moduleis the input processing module, which receives the data input 1807. Theinput processing module queues the received data, then provides it toforward error correcting module 1903. It is from the functions providedin the forward error correcting module 1903 that the FEC 1801 takes itsname.

The forward error correcting module 1903 processes the data provided bythe input processing module 1935 and then further provides it to atraining sequence and marker insertion module 1937. The trainingsequence and marker insertion module 1937 adds a training sequenceheader and may add submarkers to the data. The data is then provided toan output 1819, for acceptance by the modulator 1803.

The input processing module 1935, the forward error corrector 1903, andthe training sequence and marker insertion (TSMI) module 1935 areconnected serially to each other via asynchronous interfaces such asFIFO (First In First Out) queues. In other words, the coupling betweenthe input processing module 1935 and the forward error corrector 1903 isasynchronous, as is the connection between forward error corrector 1903and training sequence and marker insertion module (TSMI) 1937. In otherembodiments, the TSMI module may be eliminated entirely.

Data 1807 is generally coupled into the FEC 1801 via the inputprocessing module 1935. Data 1807 is provided to a multiplexer 1907which allows a microcontroller 1939 to choose between the data input1807 and a PN sequence generator 1905. The PN sequence generator 1905may be used as a data source for the FEC 1801, for example, for testing.It may also be used to provide a known signal, for example, for testingthe receiver system performance. The PN sequence module 1905 is alsocoupled to an output 1805 for use outside of the module. The data chosenby the multiplexer 1907 is queued within a first-in first-out (FIFO)queue 1909 within the input processing module 1935. Data is then furthercoupled from the FIFO 1909 to an input FIFO 1911 in the forward errorcorrecting module 1903. The forward error correcting module 1903 furtherreceives data from its input FIFO 1911 and provides it to a Reed-Solomonencoder 1913. The Reed-Solomon encoder, once it has encoded the data,then provides the Reed-Solomon encoded data to a turbo encoder 1917 andinterleaver 1915 combination.

Once the Reed-Solomon encoder 1913, the interleaver 1915, and the turboencoder 1917 completes the processing of a data symbol, it is thencoupled into an output queue 1921. The output queue 1921 is providedwith a fullness indicator. So, for example, if the FIFO 1921 is greaterthan half full, it provides an indicator signal 1931 to loop filter1929.

The FIFO 1921 further provides symbol data to the training sequence andmarker insertion TSMI module 1937. An inserter 1927 may then insert atraining sequence header and/or submarkers into the data stream, forexample, to accommodate the needs of a particular data receiver. It isthe training sequence and marker insertion module which creates theframe structure for the data. Once the training sequence and submarkershave been inserted into the data stream, the data is coupled into anoutput queue 1937. The data resides in the output queue 1937 untilcalled for by the modulator 1803. The output FIFO 1937 also has a statusindicator 1933 that indicates, for example, that it is half full. Theindicator is coupled into loop filter 1929. The FIFO level indicators1931 and 1933 may both be coupled into loop filter 1929 to determine therate at which a byte clock NCO 1811 runs.

In the described exemplary embodiment, three different clocks are usedfor control of the FEC 1801. The first clock is the byte clock NCO(Numerically Controlled Oscillator) 1811. The byte clock is used tocontrol the flow of data into the FEC 1801. The byte clock can beadjusted by the microcontroller 1939 and can also be adjusted by theoutput from the loop filter 1929, for example, based on how full FIFOs1921 and 1937 are. The forward errorcorrecting module 1903 is clocked byan FEC clock 1813. Although the FEC clock 1813 is independent of theother two clocks, it too may also be controlled by the microcontroller1939. The third clock 1815 is a bit clock. The bit clock is provided tothe TSMI module 1937 and also provided to the digital to analogconverter 2027 (as illustrated in FIG. 20). The bit clock 1815 also maybe controlled by the microcontroller 1939. The control of the clocks bythe microcontroller 1939 facilitates the ability to switch modulation ona symbol by symbol basis.

FEC Multimodulation Architecture

Because of the asynchronous nature of the modules within the FEC 1801,the flexible control of the data stream, to be modulated, can beenhanced. For example, in the described embodiment, 8PSK or QPSKmodulation can both be accommodated. Other modulation types can also beaccommodated. The modulation can be controlled from outside the FEC, forexample, utilizing the control bus 1825. Additionally, the frame lengthand header training sequence composition and placement of submarkers canbe varied on command. In addition, the turbo encoder 1917 can have avariety of coding rates, such as ⅔, ⅚, and 8/9. All the aforementionedparameters can be controlled from outside the FEC 1801. In addition, thenumber of submarkers placed within a frame can be varied in length andnumber. Submarkers may be used by the receiving system to track theincoming signal or re-establish lock. The variables can be changed on aframe-to-frame basis. For example, one frame can be modulated with 8PSKand a successive frame can be modulated with QPSK. Additionally, theturbo coding rates can be switched within successive frames. The number,size and spacing of submarkers also can be changed in successive frames.

As the above mentioned variables are changed within the FECs, variouslatencies can be encountered. The FIFOs 1909, 1911, 1921, 1923 and 1925enable the buffering of the symbol stream as parameters are changed, forexample, on a frame-to-frame or intraframe basis. The FIFOs enable thesymbol stream to be queued when parameters are changed. For example,FIFO 1911, which provides data to the turbo encoder 1917, can queueinput data as the turbo encoder is switched between rates of ⅔, ⅚ and8/9. In such a way, the input stream does not have to be started andstopped as might be the case in a synchronous type design. The incomingdata rate, however, should be controlled or it is likely that the FIFOswill either overflow or be starved for data at some point. In order toprevent the modules from being starved for data or overflowing, theoverall rate is controlled by the byte clock 1811. The byte clock iscontrolled by the loop filter which receives, as its input, FIFO statussignals 1931 and 1933. The loop filter receives signals 1931 and 1933,filters them and uses the filtered signal to control the frequency ofthe byte clock 1811. It is, of course, possible to use other FIFO statussignals from other FIFOs to control the data rate input to the modules.Which FIFOs are selected as control FIFOs depend on a variety of designvariables, for example, individual FIFO size, and peak data rates.

The feedback from FIFOs 1921 and 1925 are used to control the byte clock1811 and hence the data input rate of the data 1807. Additionallycontrol bus 1825 can communicate which variables within the FEC 1801should change. The microcontroller 1939 may not only control theparameters within the different modules of the FEC 1801, in addition italso can load the clocks, i.e., the byte clock 1811, the FEC clock 1813,and the bit clock 1815 with starting values. Tables relating the clockfrequencies to different FEC parameters can be provided in ROM (notshown) to the microcontroller 1939. When the microcontroller changesparameters within the FEC, it can also change the clock frequencies ofthe module clocks. These tables can be computed beforehand and placed inthe ROM available to the microcontroller 1939 to control the parametersused to modulate the data 1807 as it arrives.

In a second operating mode, instead of having all the clock frequenciesspecified in tabular form, the microcontroller 1939 may be providedwith, for example, a byte clock frequency. The microcontroller may thenvary the parameters within the FEC 1801 and alter the byte clock rate1811 to accommodate an estimated desired input data rate. Themicrocontroller can be provided with algorithms which examine the outputrate from the input processing module 1935 and, based on the data rateestimate, set the FEC clock 1813 to an estimated frequency. Themicrocontroller can then further examine the output rate of the forwarderror correcting module 1903, make an estimate of the bit clockfrequency that is needed, and set the bit clock to that frequency. Inother words, the microcontroller controls the parameters of the FEC 1801while the FIFOs within the modules 1935, 1903 and 1937 provide theneeded buffering to maintain a steady data flow through the FEC. Themicrocontroller can also control the clocks which process the data inthe modules 1935, 1903 and 1937 by using a table in memory, whichrelates the different parameters to the clock rate. Microcontroller 1939can also be provided with an algorithm that will predict the clock ratebased on module parameters, or the microcontroller can be provided withan algorithm or table relating the initial byte clock rate to theparameters and then examine the data flow through the module and adjustthe other clocks in line as needed. Examples of values, as may beinserted into a ROM table, are given below.

The first three examples are for a input symbol rate of 21.2 megasymbolsper second: (1) at a coding rate of ⅔, the byte clock is running at 5.13megahertz, the FEC clock is running at 42.38 megahertz, and the bitclock is running at 63.57 megahertz; (2) at a ⅚ coding rate, the byteclock is running at 6.39 megahertz, the FEC clock is running at 52.98megahertz, and the bit clock is running at 63.57 megahertz; (3) at acoding rate of 8/9, the byte clock is running at 6.81 megahertz, the FECclock is running at 56.51 megahertz, and the bit clock is running at63.57 megahertz.

The fourth through sixth examples describe clock rates versus codingrate for an input stream of 25 megasymbols per second: (4) at a coderate of ⅔, the byte clock is running at 6.05 megahertz, the FEC clock isrunning at 50 megahertz, and the bit clock is running at 75 megahertz;(5) at a coding rate of ⅚, the byte clock is running at a frequency of7.54 megahertz, the FEC clock is running at a frequency of 62.50megahertz, and the bit clock is running at a frequency of 75 megahertz;(6) at a coding rate of 8/9, the byte clock is running at 8.03megahertz, the FEC clock is running at 66.67 megahertz, and the bitblock is running at 75 megahertz.

The previous exemplary values illustrate that by knowing the inputsymbol rate and the coding rate, tables can be developed for nominalbyte clock, forward error correcting clock, and bit clock rates. Suchcomputations, as illustrated in the six examples just described, can beused to rapidly switch coding rates and clock frequencies on aframe-by-frame basis. In addition, because of the asynchronous nature ofthe modules within the FEC 1801, any temporary change in coding rate canbe buffered by the FIFOs which connect the internal processingfunctions. The overall scheme employed to keep the modules functioningwithout queue overflow or underflow is that the upstream clocks areresponsive to downstream queue sizes. Which upstream clocks areresponsive to which downstream queues are design parameters that canvary from implementation to implementation. In the described exemplaryembodiment, FIFO queues 1921 and 1925 are chosen to control the byteclock NCO 1811. Other combinations are possible such that the upstreamclock rate is controlled by a downstream FIFO size. For example, if thetraining sequence and marker insertion module 1937 were eliminated fromthe FEC 1801, as might be the case in particular embodiments, the sizeof FIFO 1921 could then alone be used to control the byte clock NCO1811.

FIG. 16 is a graphical illustration of a generalized data frame producedby the TSMI 1937, according to the present exemplary embodiment of theinvention. The data frame 2101 is, in general, composed of two differentsections. The first section in the header section as exemplified by2103. The second section is the data section as exemplified by 2105,2109 and 2113. The header is, in general, data which identifies thebeginning of a frame of data. The header may comprise various datasequences. For example, one of the data sequences within the header,according to an embodiment of the present invention, is a trainingsequence 2117. The training sequence 2117 is a known pattern of datawhich the receiver system may use to lock to the frame. The trainingsequence can be varied in length or modulation according to parametersof the satellite system. In the present exemplary embodiment, a trainingsequence size of 64 symbols is used. The remainder of the header 2103may be used for a variety of purposes, for example, but not limited to,identifying the data within the frame, identifying the modulation schemeof the data in the frame, identifying the modulation of future frames,or identifying positions of items within the data stream, such assubmarkers 2107 and 2111.

Submarkers 2107 may be inserted into the data within the frame 2101 inorder to provide a method for the receiver system to synchronize ortrack an incoming signal. In one exemplary embodiment, the submarkersize is 16 symbols, and the spacing between submarkers is 10,240symbols.

In one exemplary embodiment, the data 2105 comprises sequential blocksof turbo coded data. Submarkers may be interspersed between blocks ofdata (See FIG. 17). Such submarkers may be particularly useful when usedwith codes, such as turbo codes, which are designed to be received atlow signal levels. At low signal levels receiver systems many needmechanisms to assist in signal acquisition and training.

The submarkers may be known sequences such as, for example, copies ofthe training sequence. Such submarkers can be used to track the incomingsignal, by performing correlations on the submarker or may even be usedto acquire the signal in a case, for example, when the incoming signalis disrupted by a noise burst.

The spacing and arrangement of submarkers is further illustrated byexamples illustrated in FIGS. 17 and 18.

FIG. 17 is a graphical illustration of a frame of data having threesubmarkers. FIG. 17 comprises a training sequence 2201 followed by turboencoded blocks 1 and 2 (2203). A submarker 2205 follows turbo encodedblocks 1 and 2. Following the submarker 2205 turbo encoded blocks 3 and4 are followed by submarker 2209. Submarker 2209 is followed by turboencoded blocks 5 and 6, which are then followed by submarker 2203 whichis followed by the final turbo encoded blocks 7 and 8. Turbo encodedblocks 7 and 8 (2215) comprise the final elements within the frame. Anew frame begins immediately thereafter with training sequence 2217. Inthe example in FIG. 17, the frame comprises 41,072 symbols. The trainingsequence is 64 symbols and the submarker size is 16 symbols. The spacingbetween submarkers is 10,240 symbols.

FIG. 18 is a graphical illustration of an exemplary arrangement of datablocks and submarkers within a frame. In FIG. 18 submarkers areinterspersed within the data blocks. The frame begins with a trainingsequence (2301). One-half of turbo encoded block 1 is then positionedafter the training sequence. The half of turbo encoded block 1 2303 isfollowed by submarker 2305. Submarker 2305 is then followed by thesecond half of turbo encoded block 1 and the first half of turbo encodedblock 2 (2307). Submarker 2309 follows next, followed by the second halfof turbo encoded block 2 and the first half of turbo encoded block 3(2311). This sequence continues until the final half of turbo encodedblock 8 2335 is in place. The final half of turbo encoded block 2335marks the end of the frame. The end of the frame is followed by atraining sequence 2337 signaling the start of the next frame. Theexample illustrated in FIG. 18 includes a frame of 41,152 symbols. Thetraining sequence size is 64 symbols and the submarker size is 16symbols. The spacing between submarkers is 5,000 symbols versus 10,400in the example illustrated in FIG. 17. The spacing and submarker sizemay vary according to the characteristics of the receiver that isreceiving the signal. For example the submarkers may comprise copies ofthe training sequence.

Because of the flexibility of the system, a frame as illustrated in FIG.17 may be followed immediately by a frame as illustrated in FIG. 18. Theframe in FIG. 18 may be used, for example, when delivering data to areceiver system which is receiving a signal of low signal strength.

The following examples are not exhaustive. For example submarkers may beplaced between every block of turbo coded data, or they may be placedbetween every block of turbo coded data and additionally within theblock of turbo coded data.

The Set Top Box

A generalized block diagram of a set top box 115, according to anexemplary embodiment of the present invention, is illustrated in FIG.19. The set top box 115 receives the signal from the ODU 119. The signalreceived from the ODU 119 may include a frequency band of signals, forexample 950 to 2150 megahertz. The signal from the ODU 119 is coupledinto the exemplary set top box 115.

Within the set top box, the tuner 2901 selects the desired band offrequencies from the signal provided by the ODU 119. The desired band ofsignals is then down converted in the tuner from a high frequency signalinto a frequency which may be accepted by a signal demodulator 2903.

The demodulator 2903 accepts the down converted signal from the tuner2901, and converts the signal into a data stream. The data stream fromthe demodulator may then be provided to a turbo decoder 2905.

Within the turbo decoder 2905, the process of recreating the data sentby the transmit system is begun. As described earlier, the data streamfrom the FEC of the transmit system has been turbo code encoded to helpcorrect errors which occur in the signal. Errors in the signal occur dueto factors such as noise within the channel, atmospheric conditions,electrical interference, and a variety of other interference sources.Turbo codes are used, in part, because they provide a robust way ofencoding a signal. Turbo codes may enable a signal to be reconstructedat lower signal levels, i.e., lower signal to noise ratios, than may bepossible with other codes alone.

The turbo decoder 2905 is coupled to and cooperates with an interleaver2907. An interleaver is a device that, in the transmit system, changesthe order of the data from the order received into another order. Areceive interleaver may change the data so that it is no longer insequential order, but in a random but known sequence. A receiveinterleaver, e.g. 2907, is alternatively known as a de-interleaver. Thereceive interleaver converts the random data sequence back into normalsequential order. Interleavers may also cooperate with encoders in thetransmit system in order to encode the data signal, and decoders in thereceiver system to decode the data signal. A burst error, which affectsan adjacent sequence of bits, can be minimized if the signal has beeninterleaved. In an interleaved signal, the bits containing errors may bedecoded with other bits, which were transmitted at a time when theinterfering signal was not present. The interleaver 2907 furtherproduces a data stream which is then provided to a Reed-Solomon decoder2909.

The Reed-Solomon decoder 2909 further decodes the data stream. Bysuccessfully coding a signal with a turbo coder in conjunction with aReed-Solomon coder the signal can be made more robust, i.e., can bedecoded at lower signal to noise levels, than if either type of codinghad been used separately. The block including the demodulator 2903, theturbo decoder 2905, the interleaver 2907 and the Reed-Solomon decoder2909 are generally referred to as the receiver 2919. The output of theReed-Solomon decoder is provided to a video unit 2911. The video moduleaccepts the data from the Reed-Soloman Decoder 2909 and converts it intoa form which may be used by a user device such as the television.

The Demodulator

FIG. 20 is a block diagram of an exemplary demodulator in accordancewith an embodiment of the present invention. In-phase and quadraturesignals are received by analog to digital (A/D) converters 3103 and3105. The AID converters 3103 and 3105 are controlled by a free-runningsample clock (not shown). The frequency of the sample clock driving A/Dconverters 3105 and 3103 is a non-integer multiple of the symbol rate.

An automatic gain control (AGC) 3107 examines the signal size at theoutput of the A/D converters 3103 and 3105. Maximum resolution can beobtained with a input signal whose maximum size corresponds to the rangeof the A/D converters. The AGC controls the gain of amplifiers thatamplify input signals prior to providing them to the A/D converters.From the A/D converters the data is provided to the multiplier block3113. In the multiplier block a digital frequency shift is introduced tocancel any frequency offset which exists between the clocks of thetransmitter system and receiver system.

The frequency loop 3115 serves to change the multiplication factor andthereby cancel the frequency offset. The data is then coupled intodecimation filters 3119 and 3121. The decimation filters reduce the datarate, thereby filtering the data. From the decimation filters the signalvalues are coupled into symbol sample (SS) blocks 3127 and 3129. The SSblocks 3127 and 3129 extract symbol values from the signal valuespresented to it by the decimation filter. The symbol extraction issynchronized using a symbol timing loop 3131. The symbols retrieved bythe SS blocks are next provided to Nyquist filters 3133 and 3135, whichprovide the symbols with Nyquist filtering. From the Nyquist filter thesymbols are coupled to the demodulator output 3109 and 3111.

Receiver Synchronization with Frequency Offset

As explained earlier, in a receiver system, such as illustrated in FIG.20, two basic problems need to be dealt with. The first problem is thatthe center frequency of the signal, which the receiver system isattempting to synchronize with, may be offset by a considerable amount(referred to as the delta frequency). The delta frequency arises becausethe receiver clock is not the same frequency as the transmitter clock.The receiver system must adjust its clock to match the transmissionclock. The frequency of the transmission clock, also called the carrierfrequency, may also vary over time. A second problem is that transients,such as intermittent noise, changing atmospheric conditions, andinterference from other signal sources, may cause relatively shortduration deviations or interruptions of the signals input to the system.These deviations may cause transient phase deviations, which need to beaccounted for by the system. The process of matching the carrierfrequency to the receiver frequency is often referred to as acquisition,and the process of maintaining phase lock to a signal, despite transientnoise and interference conditions, is often called tracking.

The frequency offset of the incoming signal is adjusted within themultiplier 3113, illustratively a digital complex multiplier. The basicoperation of the multiplier 3113 is to multiply the incoming signal andthereby create a frequency translation of the incoming frequency. Themultiplier 3113 will create frequency images equal to the inputfrequency plus the multiplying frequency and also equal to inputfrequency minus the multiplying frequency. The multiplying frequency isprovided from within a frequency loop block 3115.

The frequency loop 3115 includes a phase detector 3149, which detectsthe phase difference of the outputs of the Nyquist filters 3133 and 3135and the input signal as provided by multiplier 3113.

The symbol timing loop 3131 adjusts the timing of symbol sampling, thatis it selects the value which will be decoded as a symbol.

When the system is initially turned on, both the frequency loop 3115 andthe symbol timing loop 3131 are disabled by the correlator 3101. Thecorrelator then examines the incoming symbols and estimates, thefrequency offset between the receiver clock and the transmitter clock,as previously described in the section on the differential correlator.Once the correlator 3101 has estimated the difference between thecarrier frequency and the receiver clock frequency, the value of theoffset between the transmitter frequency and the receiver frequency,that value can be inserted, as a starting value, into the frequency loop3115. Once the offset frequency has been loaded into the frequency loopby the correlator, the correlator may then enable the loop. Thefrequency loop can then quickly lock in and track the carrier frequency.The header processor also estimates the phase offset within the symboltiming loop 3131. Once the correlator-processor 3101 has an estimate ofthe phase offset in the symbol timing loop, it can be inserted into thesymbol timing loop. When the header processor couples the initial valueinto the symbol timing loop 3131, the header processor may enable thesymbol timing loop 3131. By providing starting value for the frequencyloop 3115, the acquisition time of the frequency loop 3115 is decreasedover a conventional type methodology, in which the phase detector alonecauses the synchronization of the loop starting at an arbitrary value.Similarly, by providing an initial value for the symbol timing loop 3131and then enabling the loop, the tracking function within the symboltiming loop can lock to the symbol stream much faster than if it hadstarted from an arbitrary value. Additionally, because the frequency ofthe loop 3115 does not traverse a range of values, the false lockproblem, in which the system may lock to an incorrect frequency, isminimized.

Receiver Synchronization using Interspersed Data

Additionally, in order to aid acquisition and tracking, thecorrelator-processor can enable the symbol timing loop 3131 in such amanner that only the training sequence and submarkers are used forsynchronizing the symbol timing loop 3131. The same training sequenceand submarkers may also be gated into the frequency timing loop 3115 forthe purpose of synchronizing the frequency timing loop 3115.

Although a preferred embodiment of the present invention has beendescribed, it should not be construed to limit the scope of the appendedclaims. Those skilled in the art will understand that variousmodifications may be made to the described embodiment. Moreover, tothose skilled in the various arts, the invention itself herein willsuggest solutions to other tasks and adaptations for other applications.It is therefore desired that the present embodiments be considered inall respects as illustrative and not restrictive, reference being madeto the appended claims rather than the foregoing description to indicatethe scope of the invention.

1. A transmitter comprising: a turbo encoder for generating a pluralityof turbo encoded data blocks; an inserter for generating a data frame byinserting a training sequence before the turbo encoded data blocks; andan output port for transmitting the data frame.
 2. The transmitter ofclaim 1 further comprising a modulator coupled to the inserter formodulating the data frame before the data frame is transmitted.
 3. Thetransmitter of claim 1 further comprising a submarker insertion devicecoupled to the turbo decoder to insert a plurality of submarkers withinthe turbo encoded data blocks.
 4. The transmitter of claim 1 furthercomprising an input adapted to receive a plurality of turbo encoded datablocks from the turbo encoder including an input queue.
 5. Thetransmitter of claim 4 wherein the input queue comprises afirst-in-first-out storage device.
 6. The transmitter of claim 2 furthercomprising a forward error correction device including an output adaptedto buffer the data frame, the output being coupled to the modulator. 7.The transmitter of claim 3 wherein the submarker insertion device isadapted to insert one of the submarkers between two turbo encoded datablocks.
 8. The transmitter of claim 3 wherein the submarker insertiondevice is adapted to insert each of the submarkers between differentpairs of the turbo encoded data blocks.
 9. The transmitter of claim 3wherein the submarker insertion device is adapted to insert one of thesubmarkers within one of the turbo encoded data blocks.
 10. Thetransmitter claim 3 wherein the submarker insertion device is adapted toinsert each of the submarkers within a different one of the turboencoded data blocks.
 11. The transmitter of claim 3 wherein thesubmarker insertion device is programmable as to the insertion of thesubmarkers within the turbo encoded data blocks.
 12. The transmitter ofclaim 3 wherein the submarker insertion device is programmable to insertthe submarkers between the turbo encoded data blocks or insert thesubmarkers within the turbo encoded data blocks.
 13. The transmitter ofclaim 6 wherein the forward error correction further comprises an inputencoder, and an interleaver disposed between the input encoder and theturbo encoder.
 14. The transmitter of claim 13 wherein the input encodercomprises a Reed Solomon encoder.
 15. A transmitter comprising: a turboencoder for generating a plurality of turbo encoded data blocks; asubmarker insertion device coupled to the turbo encoder for inserting atleast one submarker within the turbo encoded data blocks; and aninserter for creating a data frame by inserting a training sequencebefore the turbo encoded data blocks.
 16. The transmitter of claim 15further comprising a modulator coupled to the inserter for modulatingthe data frame before the data frame is transmitted.
 17. The transmitterof claim 16 further comprising a forward error correction deviceincluding an output adapted to buffer the data frame, the output beingcoupled to the modulator.
 18. A method for transmitting data, the methodcomprising: generating a plurality of turbo encoded data blocks;generating a data frame by inserting a training sequence before theturbo encoded data blocks; and transmitting the data frame.
 19. Themethod of claim 18 further comprising modulating the data frame beforetransmitting the data frame.
 20. The method of claim 18 furthercomprising inserting at least one submarker within the turbo encodeddata blocks.